Electronics Guide

Every electronic device is an unintentional radiator, emitting electromagnetic energy that can interfere with other equipment, disrupt wireless communications, and violate regulatory standards. Understanding the specific sources of these emissions is the first step toward controlling them. While electromagnetic compatibility (EMC) engineers often focus on shielding and filtering as remediation strategies, the most effective approach addresses emissions at their origin.

Radiated emissions arise from multiple mechanisms within electronic systems. Clock signals and their harmonics, differential-mode currents flowing in signal loops, common-mode currents on cables, printed circuit board structures, enclosure apertures, connectors, heat sinks, individual components, and switching power supplies all contribute to the overall emission profile. Each source has distinct characteristics, frequency ranges, and mitigation strategies. This comprehensive examination of emission sources provides the foundation for developing effective EMC design practices.

Clock Harmonics Radiation

Clock signals represent the most predictable and often the most problematic sources of radiated emissions in digital systems. Unlike random data signals, clocks are continuous, periodic waveforms that concentrate their energy at discrete frequencies: the fundamental clock frequency and its harmonics. This spectral concentration produces narrow, high-amplitude peaks that readily exceed regulatory emission limits.

Harmonic Content of Clock Signals

An ideal square wave contains only odd harmonics, with amplitudes decreasing as 1/n where n is the harmonic number. A 50 MHz clock thus contains energy at 50 MHz, 150 MHz, 250 MHz, 350 MHz, and so on. The third harmonic is at one-third the fundamental amplitude (approximately -10 dB), the fifth harmonic at one-fifth (-14 dB), and so forth. However, real clock signals are trapezoidal rather than perfectly square, with finite rise and fall times that modify this harmonic structure.

The rise and fall times of a clock signal determine the bandwidth of its harmonic content. A signal with rise time t_r has significant harmonic content up to approximately f = 0.35/t_r. For a clock with 1 ns rise time, harmonics extend to roughly 350 MHz with significant amplitude. With 500 ps rise times typical in modern high-speed logic, harmonic content extends to 700 MHz and beyond. These high-order harmonics are particularly problematic because they coincide with frequencies where PCB traces and cables become efficient antennas.

The spectral envelope of a trapezoidal waveform follows a characteristic shape: flat from DC to the corner frequency f₁ = 1/(pi times the pulse width), then decreasing at 20 dB per decade until frequency f₂ = 1/(pi times the rise time), after which it decreases at 40 dB per decade. Understanding this envelope helps predict which harmonics will dominate emissions and guides the selection of edge rate limiting strategies.

Multiple Clock Domains

Modern digital systems often contain multiple clock signals at different frequencies: processor clocks, memory clocks, peripheral interface clocks, and communication clocks. Each clock generates its own harmonic series, creating a dense spectrum of potential emission frequencies. Intermodulation between clocks can generate additional spurious frequencies that complicate troubleshooting.

When clock frequencies are harmonically related (such as a 100 MHz clock and its 25 MHz divided output), their harmonic series share common frequencies, potentially reinforcing emissions at those points. Conversely, unrelated clock frequencies produce independent harmonic series that may spread emission energy across more frequencies, reducing peak amplitudes at any single frequency.

Mitigation Strategies for Clock Emissions

Controlling clock harmonic radiation begins at the source. Use the slowest clock frequency that meets functional requirements, and specify the slowest acceptable edge rates. Many clock drivers offer selectable output impedance or slew rate control, allowing optimization for EMC rather than maximum speed. Series resistors at clock outputs provide simple edge rate control, increasing rise and fall times to reduce high-frequency harmonic content.

Spread-spectrum clocking (SSC) modulates the clock frequency over a small range, typically 0.5% to 2% of the center frequency. This spreads the harmonic energy across a bandwidth rather than concentrating it at discrete frequencies. Peak emission levels at any specific frequency decrease by 10-20 dB depending on modulation parameters, often making the difference between passing and failing EMC compliance tests. SSC is particularly effective for clock harmonics because they have the narrowest spectral peaks.

Proper clock distribution design minimizes radiation from clock traces. Keep clock traces short and route them on internal PCB layers between ground planes. Use differential clock signaling for high-speed clocks, which reduces radiation through field cancellation. Terminate clock lines properly to prevent reflections that increase emission levels. Buffer clocks locally rather than distributing from a central source, reducing trace lengths and the number of clock traces crossing the board.

Differential-Mode Radiation Mechanisms

Differential-mode radiation occurs when current flows in a closed loop formed by a signal conductor and its return path. This current loop acts as a magnetic dipole antenna, radiating electromagnetic energy proportional to the loop area, the current amplitude, and the square of the frequency. Even small loops carrying modest currents can produce significant radiation at high frequencies.

The Loop Area Problem

The electric field radiated by a small current loop is given by E = 1.316 times 10^-14 times f² times A times I / r, where E is the electric field in V/m, f is the frequency in Hz, A is the loop area in m², I is the current in amperes, and r is the distance from the source in meters. The f² dependence means that doubling the frequency quadruples the radiated field, explaining why high-frequency signals and fast digital edges create the most severe radiation problems.

Consider a 10 cm by 1 cm loop (100 mm² or 10^-4 m²) carrying 10 mA of current at 100 MHz. At 3 meters distance, this produces an electric field of approximately 14 microvolts per meter, or about 23 dB(microvolts/meter). This single small loop could approach or exceed Class B emission limits at 100 MHz. At 300 MHz (the third harmonic of a 100 MHz clock), the same loop produces a field nine times larger, easily exceeding limits.

Sources of Current Loops

Current loops form whenever signal current takes a path separated from its return current. On a properly designed PCB with continuous ground planes, signal return current flows in the ground plane directly beneath the signal trace, creating a minimal loop area equal to the trace width times the dielectric thickness. This is why solid ground planes are fundamental to low-emission design.

Loop area increases dramatically when ground plane continuity is disrupted. Slots in ground planes force return current to flow around the slot, increasing the effective loop area by orders of magnitude. A 5 cm slot in a ground plane beneath a signal trace can increase loop area from a few square millimeters to tens of square centimeters, transforming a low-emission design into a severe EMC problem.

Connector transitions often create large loops. When signals transition from one PCB layer to another, return current must also change layers. Without adequate via stitching near signal vias, return current flows through long paths, creating large loops. Similarly, signals crossing from one board to another through connectors can create loops if ground return paths are not provided in close proximity to signal paths.

Reducing Differential-Mode Radiation

Minimizing differential-mode radiation requires systematic attention to loop area throughout the design. Maintain continuous ground planes with no slots or splits beneath high-speed signal routes. When ground plane breaks are necessary for other reasons, route high-speed signals away from these areas or provide ground stitching capacitors across the gap.

For signals changing layers, place ground return vias immediately adjacent to signal vias. A ground via within 1-2 mm of a signal via reduces loop area by an order of magnitude compared to relying on distant ground connections. For critical high-speed signals, use multiple return vias around the signal via to further reduce inductance and loop area.

Differential signaling inherently minimizes loop area because the two signal lines carry equal and opposite currents in close proximity. The fields from these opposed currents largely cancel in the far field, reducing radiation significantly compared to single-ended signals with the same edge rates and frequencies. Use differential signaling for high-speed clocks, data buses, and any signals with fast transitions.

Common-Mode Radiation from Cables

Common-mode radiation typically dominates the radiated emissions from electronic products, particularly those with external cables. Common-mode currents flow in the same direction on all conductors, using the chassis, ground plane, or earth as the return path. Even microampere levels of common-mode current on cables can produce radiated fields exceeding regulatory limits because the entire cable acts as an efficient antenna.

Cable Antenna Behavior

External cables function as monopole antennas when carrying common-mode current. A cable of length L becomes resonant at frequencies where L equals odd multiples of quarter wavelengths, with the first resonance at f = c/(4L) where c is the speed of light. A 1-meter cable resonates at approximately 75 MHz, a 2-meter cable at 37.5 MHz. At resonance, radiation efficiency peaks and even small common-mode currents produce large radiated fields.

Above the first resonance, cables remain efficient radiators at all higher frequencies, with radiation efficiency increasing with frequency. This broadband antenna behavior means that common-mode currents at any frequency above about 30 MHz (typical lower limit for radiated emission testing) can create significant emissions. Cable radiation is often the limiting factor for EMC compliance.

Origins of Common-Mode Currents

Common-mode currents arise from multiple mechanisms. Ground potential differences between circuit boards or between a board and the chassis create driving voltages that force common-mode current onto cables. In systems with separate power and signal grounds, even millivolt potential differences can drive problematic common-mode currents.

Asymmetries in differential circuits generate common-mode currents. A differential driver with slightly unequal rise and fall times, or a differential pair with length mismatch, converts some differential signal energy to common mode. Driver skew of just 100 ps can create sufficient common-mode current to cause emission problems.

Capacitive coupling from internal circuits to cables and chassis provides another common-mode current path. High-voltage switching nodes, particularly in power supplies, can couple noise to nearby cables through parasitic capacitance. Even careful layout cannot eliminate all parasitic coupling paths in complex systems.

Quantifying Common-Mode Emissions

The relationship between common-mode current and radiated field strength is approximately E = 1.257 times 10^-6 times f times I times L / r for a short monopole (L less than quarter wavelength), where E is in V/m, f in Hz, I in amperes, L in meters, and r in meters. For a 1-meter cable carrying 10 microamperes at 100 MHz measured at 3 meters: E equals approximately 42 microvolts per meter, or about 32 dB(microvolts/meter), potentially exceeding Class B limits.

This calculation illustrates why common-mode currents are so problematic: microampere currents that are negligible from a circuit function perspective can dominate radiated emissions. The long cable length (acting as antenna size) and the linear frequency dependence (compared to f² for differential mode) make common-mode radiation efficient even at moderate frequencies.

Controlling Common-Mode Radiation

The most effective common-mode control prevents common-mode currents from reaching cables in the first place. Use common-mode chokes at cable entry points to present high impedance to common-mode currents while passing differential signals unimpeded. Ferrite cores placed over cables provide simple, inexpensive common-mode suppression. For severe problems, use split ferrites or wound common-mode chokes for higher impedance.

Filter connectors integrate capacitors that shunt high-frequency common-mode currents to chassis ground at the enclosure boundary. This approach is particularly effective because it grounds noise before it reaches the cable. For maximum effectiveness, the connector must bond directly to the chassis with low impedance around its entire periphery.

Ground system design affects common-mode current generation. Single-point grounding reduces ground loops but can create high-impedance ground paths that allow ground potential differences. Multi-point grounding reduces potential differences but can create ground loops. The optimal approach depends on system size and frequency range, often requiring hybrid grounding strategies.

Cable shield grounding strategy impacts common-mode radiation. For low frequencies, grounding the shield at one end minimizes ground loop currents but allows common-mode current to flow on the shield exterior. For high frequencies, grounding at both ends provides lower transfer impedance but can create ground loops. Use 360-degree shield terminations to connectors for best performance, avoiding pigtail connections that create inductance and defeat shield effectiveness.

Printed Circuit Board Radiation

Printed circuit boards contain numerous structures that can radiate electromagnetic energy. Traces act as transmission lines and antennas, power distribution networks carry high-frequency switching currents, and the board edges can radiate from fringing fields. Understanding PCB radiation mechanisms enables layout practices that minimize emissions.

Trace Radiation Mechanisms

A PCB trace above a ground plane forms a transmission line with characteristic impedance determined by trace geometry. When properly terminated, signal energy propagates along the line without radiation. However, several conditions can cause traces to radiate.

Unterminated or improperly terminated traces create reflections that can cause standing waves. At frequencies where the trace length approaches odd multiples of quarter wavelengths, resonance occurs and radiation efficiency increases dramatically. A 7.5 cm trace resonates at approximately 1 GHz (quarter wave in FR-4 with effective dielectric constant around 4).

Trace discontinuities cause radiation. Any change in trace geometry, including bends, vias, width changes, and splits, creates impedance discontinuities that partially reflect signals and radiate energy. Sharp 90-degree bends are particularly problematic at high frequencies, where they act as small loop antennas. Use 45-degree bends or curved traces for high-speed signals.

Traces routed near board edges radiate more efficiently than traces over continuous ground planes. Edge fields extend into free space rather than terminating on the ground plane, creating a partially unshielded transmission line. Keep high-speed signals at least 3-5 times the trace height from board edges.

Power Distribution Radiation

Power distribution networks carry AC currents at frequencies from DC through the highest switching frequencies in the system. These currents flow between power and ground planes, through vias, and across plane boundaries. Non-ideal current paths create loops that radiate.

Power plane resonances occur when the plane dimensions match electrical wavelengths. At resonance, standing waves develop on the planes, creating localized high-field regions that can couple to other circuits or radiate from board edges. Power plane resonances typically occur above 100 MHz for common board sizes, with frequencies determined by plane dimensions and dielectric properties.

Decoupling capacitors interrupt the current path between power and ground planes at each IC, storing local energy for high-frequency demands. However, capacitors have inductance that limits their effectiveness at high frequencies. The resonant frequency f = 1/(2pi times square root of LC) determines the useful frequency range of each capacitor. Above this frequency, capacitor impedance increases and effectiveness decreases.

Board Edge Radiation

PCB edges can radiate from the fringing fields between power and ground planes. At high frequencies, these fringing fields extend into free space and can couple to the environment or to nearby cables. The effect is most pronounced when high-frequency currents flow in the planes or when plane resonances create high field regions near edges.

Edge radiation reduction techniques include shorting the power and ground planes together with vias along board edges (stitching vias), using buried power planes away from board edges, and placing ground traces along board edges to contain fringing fields. Some designs add metallic edge shields that connect to ground to provide additional containment.

Enclosure Leakage and Slots

Metal enclosures provide the primary defense against radiated emissions, containing electromagnetic fields within the product. However, real enclosures contain numerous apertures that compromise shielding effectiveness. Understanding aperture radiation mechanisms enables effective enclosure design.

Slot Antenna Behavior

A slot in a conductive surface acts as an antenna, with radiation characteristics complementary to those of a wire dipole of the same dimensions. The slot becomes resonant when its length approaches half a wavelength, at which point radiation efficiency peaks. A 10 cm slot resonates at approximately 1.5 GHz in free space (somewhat lower on a real enclosure due to edge effects).

However, slots need not be resonant to radiate significantly. When driven by internal electromagnetic fields, even sub-resonant slots allow field leakage that can cause emission failures. The shielding effectiveness reduction depends on the ratio of slot length to wavelength and the field strength at the slot location.

Common Sources of Slots

Panel seams form long slots when mating surfaces do not make continuous electrical contact. A seam running along a 30 cm edge can resonate at 500 MHz and radiate at lower frequencies as well. Painted or anodized surfaces create non-conductive barriers that prevent metal-to-metal contact, effectively creating slot apertures.

Ventilation openings are intentional apertures that can dominate enclosure radiation. Large rectangular openings are particularly problematic because they behave as slot antennas. Circular holes have better shielding properties for equal area because their maximum dimension (diameter) is smaller than the diagonal of an equivalent-area square.

Display windows and control panel cutouts create large apertures in enclosure surfaces. These openings may be located near high-field regions inside the enclosure, making them efficient radiators. Touch screen implementations add complexity because the touch sensor must remain electrically accessible while maintaining EMC performance.

Seam Treatment for Low Leakage

Creating low-impedance seams requires continuous electrical contact along the entire seam length. For bare metal surfaces, sufficient fastener pressure and surface flatness can achieve good contact. Fastener spacing should be less than one-twentieth wavelength at the highest frequency of concern: for 1 GHz, this means less than 15 mm spacing.

When painted or finished surfaces prevent direct metal contact, conductive gaskets maintain electrical continuity. Beryllium copper finger stock, wire mesh gaskets, and conductive elastomer gaskets all provide seam bonding. The gasket must compress sufficiently to make reliable contact with both surfaces while accommodating manufacturing tolerances.

Some designs use conductive coatings on mating surfaces, removing paint only in the gasket contact area. Others specify conductive anodize or chromate conversion coatings that provide both corrosion protection and electrical conductivity. These approaches maintain enclosure appearance while achieving EMC performance.

Ventilation Aperture Design

Ventilation requirements often conflict with shielding requirements. Designing apertures that satisfy both requires understanding the relationship between aperture size, shape, and shielding effectiveness.

Multiple small holes provide better shielding than fewer large holes of equal total area. For a given total open area, the radiated field decreases approximately as the square of the number of holes. Thus, 100 holes of 3 mm diameter provide much better shielding than 10 holes of 9.5 mm diameter, despite equal total area.

Honeycomb ventilation panels provide excellent shielding while maintaining airflow. The honeycomb cells act as waveguides below cutoff, attenuating electromagnetic energy while passing air. Cell dimensions determine the cutoff frequency, above which shielding effectiveness degrades. Standard honeycomb panels provide 40-80 dB of shielding through several GHz.

Connector and I/O Radiation

Connectors represent critical points where signals cross the enclosure boundary, bringing together cables, apertures, and filtering requirements. Poor connector design or implementation can dominate system emissions even when internal circuits and external cables are well-controlled.

Connector Aperture Effects

Every connector creates an aperture in the enclosure. The connector cutout allows fields to leak from inside to outside the enclosure, and the interface between connector and enclosure may not provide continuous shielding. Standard commercial connectors often have gaps between the connector shell and the enclosure that create slot antennas.

D-subminiature connectors, common in legacy designs, have particular problems. The connector shell makes limited contact with the enclosure, typically only through the mounting screws. The gap between shell and enclosure can be several millimeters, creating an effective slot antenna. The pins themselves pass through without filtering, carrying both intentional signals and unintentional noise.

Connector Shell Grounding

Proper connector grounding requires low-impedance bonding between the connector shell and the enclosure around the entire connector periphery. Backshell connectors with 360-degree contact to the enclosure provide this capability. The backshell clamps around the cable shield and bonds to the enclosure, maintaining shielding continuity from cable through connector into enclosure.

Panel-mount connectors should have flanges that make continuous contact with the enclosure around the cutout. Gaskets between the flange and enclosure maintain contact even with surface finish variations. Some connectors include integral EMI gaskets for this purpose.

Avoid grounding connector shells through wires or pigtails, which add inductance and create loop areas. A 25 mm pigtail connection has approximately 25 nH of inductance, presenting nearly 16 ohms of impedance at 100 MHz. This high-impedance path cannot effectively drain high-frequency currents and may actually increase radiation by creating a resonant structure.

Filter Connectors

Filter connectors integrate capacitive filtering directly into the connector structure, providing bulkhead-feedthrough capacitors on each pin. This arrangement shunts high-frequency noise to chassis ground at the enclosure boundary, preventing noise from reaching external cables or entering from external sources.

The filter capacitors typically range from 100 pF to 10 nF depending on the intended signal bandwidth and attenuation requirements. Higher capacitance provides more attenuation but limits signal bandwidth. For digital signals, capacitance must be low enough to pass the intentional signal frequencies without excessive distortion.

Pi-filter connectors add inductance in series with the signal pins in addition to shunt capacitance, providing steeper attenuation versus frequency. These are used for severe EMI environments or when high attenuation is required without excessive capacitor loading.

I/O Circuit Design

The circuits driving I/O connectors significantly affect radiation. Fast-edge signals driving long cables radiate more than slow-edge signals. Include slew rate limiting on all outputs that do not require maximum speed. Series resistors at IC outputs slow edges and reduce high-frequency content.

Provide common-mode filtering at the connector, typically with ferrite beads or common-mode chokes. These components add impedance in the common-mode path while minimally affecting differential signals. Place filtering as close as possible to the connector to minimize the length of unfiltered trace between the filter and the cable.

ESD protection devices at I/O pins also affect EMC. Transient voltage suppressors (TVS) and ESD diodes can conduct during fast signal transitions, creating additional paths for common-mode current. Select ESD protection devices with attention to their capacitance and behavior at signal frequencies, not just their protection characteristics.

Heat Sink Antenna Effects

Heat sinks are essential for thermal management in power electronics and high-performance processors, but they can also function as efficient antennas when coupled to switching circuits. The physical dimensions of heat sinks, often several centimeters in each direction, are comparable to wavelengths at frequencies commonly found in switching power supplies and processor clocks.

Coupling Mechanisms

Heat sinks mount in intimate contact with switching devices, creating capacitive coupling between the device and the heat sink. Even with insulating thermal pads, the capacitance between a power transistor die and a heat sink can be 10-50 pF. At 100 MHz, 10 pF represents approximately 160 ohms, easily low enough to couple significant current into the heat sink.

When the switching device operates, high-frequency currents flow into the heat sink through this parasitic capacitance. The heat sink then radiates as a monopole or dipole antenna depending on its mounting and grounding configuration. Because heat sinks are often exposed on enclosure exteriors or thermally connected to chassis, they have efficient coupling to the external environment.

Heat Sink Resonances

Heat sink fins create complex resonant structures. Individual fins can resonate as quarter-wave monopoles, and combinations of fins form resonant arrays. A 5 cm fin resonates at approximately 1.5 GHz, while the overall heat sink dimensions may create resonances in the 100-500 MHz range typical of switching power supply harmonics.

When switching frequencies or their harmonics coincide with heat sink resonances, radiation increases dramatically. This explains why some products have emission peaks at seemingly random frequencies that do not correlate with obvious clock or switching frequencies: the emissions occur where harmonic frequencies coincide with mechanical resonances.

Mitigation Approaches

Grounding the heat sink provides a return path for coupled currents, reducing radiation. Connect heat sinks to ground with low-impedance paths, preferably multiple connections around the perimeter. If the heat sink must be electrically isolated from ground for safety or circuit reasons, use RF grounding through capacitors that pass high frequencies while blocking DC.

Reduce coupling between switching devices and heat sinks by using thicker insulating materials or lower-capacitance thermal interface materials. Faraday shields between the device and heat sink intercept coupled fields and shunt them to ground, preventing heat sink excitation. These shields must be grounded with very low impedance to be effective.

For severe problems, enclose the heat sink within a shielded compartment inside the main enclosure. This adds weight, cost, and thermal design complexity but provides definitive containment of heat sink radiation. Air channels or thermal conduits conduct heat through the shield to external cooling surfaces.

Component-Level Emissions

Individual electronic components generate electromagnetic emissions that can contribute to system-level radiated emissions. While component-level emissions are generally smaller than system-level emissions from cables and enclosure apertures, they can dominate in well-designed systems where other sources are controlled.

Integrated Circuit Emissions

Integrated circuits emit electromagnetic energy from multiple mechanisms: power supply current transients, internal clock distribution, I/O switching, and package lead inductance. Modern high-speed processors with clock frequencies of several GHz and millions of transistors switching simultaneously generate significant broadband emissions.

IC package design affects emissions. Bond wire inductance creates voltage drops during current transients, coupling noise to package leads. Flip-chip packages with shorter interconnections produce lower emissions than wire-bonded packages. Package shielding, available on some RF and precision analog ICs, contains emissions within the package.

The relationship between IC technology and emissions is complex. Smaller technology nodes operate at lower voltages, reducing the energy per switching event, but they operate at higher speeds with more transistors, increasing the number of events per second. Generally, each technology generation reduces per-transistor emissions while increasing per-chip emissions.

Discrete Component Emissions

Passive components generally do not directly generate emissions, but they can radiate when carrying high-frequency currents. Inductors with ferrite cores can radiate from incomplete magnetic field containment. Capacitors with high equivalent series resistance (ESR) dissipate energy and can develop significant voltage drops during current transients.

Switching semiconductors (transistors, MOSFETs, IGBTs) generate emissions during switching transitions. Fast switching reduces power dissipation but increases high-frequency emission content. The switching waveform, particularly the ringing that often follows transitions, contains significant high-frequency energy that can radiate from associated circuit structures.

Crystal Oscillators

Crystal oscillators are potential emission sources because they generate precise, stable frequencies that concentrate energy at specific points in the spectrum. The oscillator itself typically radiates minimally due to its small size, but it drives other circuits that can radiate. Crystal oscillator output traces should be treated as emission-critical signals.

Some oscillator packages include internal output buffers that add significant drive capability but also increase emissions. Lower-drive oscillators produce less current available to flow in trace antennas but may require additional buffering, which should be done with controlled-edge-rate buffers.

Measuring Component Emissions

Near-field probes enable measurement of emissions from individual components. Small loop probes detect magnetic field emissions from current-carrying structures, while electric field probes detect voltage-related fields. By positioning probes close to components, their individual contributions to overall emissions can be assessed.

Comparison between similar components from different manufacturers can reveal significant emission differences. A component with 10 dB lower emissions might enable system compliance without additional filtering or shielding. Component selection based on EMC performance, in addition to traditional parameters, is an increasingly important design consideration.

Switching Power Supply Radiation

Switching power supplies are among the most significant sources of radiated emissions in electronic products. Their operation involves high-voltage, high-current switching at frequencies typically between 50 kHz and several MHz, with fast edges that generate harmonics extending into hundreds of MHz. Every switching power supply is an intentional generator of high-frequency energy that must be carefully contained.

Switching Waveform Emissions

The fundamental switching frequency and its harmonics extend across a wide frequency range. A 100 kHz switcher with 50 ns switching edges generates harmonics with significant amplitude through at least 10 MHz (1/pi times 50 ns), with energy detectable at even higher frequencies. These harmonics can coincide with regulatory measurement frequencies and cause compliance failures.

Switching transients are particularly problematic. The fast voltage and current transitions during switching create broadband emissions. Switch node ringing, caused by parasitic inductance and capacitance in the switching loop, produces damped oscillations at frequencies determined by these parasitics, often in the 10-100 MHz range where many radiated emission limits apply.

Common-Mode Noise Generation

Switching power supplies are prolific generators of common-mode noise. The high dV/dt at switching nodes couples capacitively to heat sinks, transformers, enclosures, and cables. Primary-to-secondary transformer capacitance provides a path for common-mode current between isolated circuit sections. Output cables carry common-mode current generated by these mechanisms.

Input cables also carry common-mode current injected by the power supply. Line filter design must address both differential-mode noise (conducted between line and neutral) and common-mode noise (conducted between both line conductors and ground). Common-mode noise is often more difficult to filter because the entire input cable becomes a radiating antenna.

Transformer and Inductor Radiation

Magnetic components in switching power supplies can radiate directly. Transformer leakage flux and inductor fringing fields extend into the surrounding space. These fields decrease rapidly with distance but can couple to nearby circuits or enclosure structures. Shielded magnetic components contain these fields but add cost and may have reduced electrical performance.

Transformer interwinding capacitance provides a path for common-mode current between primary and secondary. Faraday shields between windings intercept this capacitive coupling and shunt the coupled current to ground, but the shield must be properly connected and must not create its own problems as a resonant structure.

Design Techniques for Low Emissions

Minimize switching loop area to reduce magnetic field radiation. Place the input capacitor, switch, transformer (or inductor), and output diode in the tightest possible loop. Use low-inductance capacitors and connections. Place damping snubbers to control ringing without excessive power dissipation.

Control switching edges to the extent that efficiency allows. Slower switching reduces high-frequency harmonic content but increases switching losses. The optimal balance depends on the operating frequency and the frequency range of emission concern. Gate resistors on power MOSFETs provide simple edge rate control.

Shield the power supply section from other circuits and from the external environment. A power supply compartment within the main enclosure contains radiated fields. Input and output filtering at the compartment boundary prevents conducted emissions from reaching external cables.

Spread-spectrum modulation of the switching frequency can reduce peak emissions by spreading harmonic energy. This technique is increasingly common in commercial power supply controllers. The switching frequency varies over a small percentage range, typically 5-10%, reducing peak emissions at any single frequency by 6-12 dB.

Identifying and Diagnosing Emission Sources

When radiated emission testing reveals failures, identifying the specific sources is essential for developing effective corrective actions. Systematic troubleshooting isolates each potential source and quantifies its contribution to overall emissions.

Frequency Domain Analysis

Examine the emission spectrum for clues about the source. Emissions at exact multiples of a known clock frequency clearly originate from that clock signal or circuits driven by it. Emissions at odd-looking frequencies may result from intermodulation between multiple sources or from resonances in mechanical structures.

The emission shape provides information. Sharp, narrow peaks indicate coherent sources like clocks. Broader peaks or pedestals suggest resonances being excited by broadband noise. Broadband elevation across wide frequency ranges indicates wideband sources like fast switching edges or digital data.

Spatial Localization

Near-field probing identifies the physical location of emission sources. Scan the PCB, cables, and enclosure with electric and magnetic field probes while monitoring the frequency of interest. Higher field readings indicate proximity to the source or to an efficient radiating structure.

Note that near-field maximum locations may not correspond to far-field emission sources. A small high-field region may radiate less than a larger moderate-field region. Near-field scanning identifies candidate areas for attention; verification requires far-field measurements or modeling to confirm which candidates actually contribute to measured emissions.

Configuration Variation

Systematically varying the product configuration helps isolate emission sources. Disconnect cables one at a time while monitoring emissions. If emissions decrease when a particular cable is disconnected, that cable is involved in the radiation mechanism, either as the antenna or as a path for common-mode current.

Change operating modes to identify which circuits are active when emissions are highest. If emissions disappear when a particular function is disabled, the circuits associated with that function are likely sources. This approach is particularly useful for identifying software-dependent emissions that appear only in certain operating states.

Temporary Modifications

Temporary modifications test hypotheses about emission sources and mitigation approaches. Adding ferrite cores to cables tests whether common-mode cable radiation is significant. Covering apertures with conductive tape tests whether enclosure leakage contributes. Adding temporary shielding around circuit areas tests whether those areas are radiating.

These temporary modifications need not be production-ready to provide useful information. If a modification reduces emissions, a production-appropriate implementation of the same approach will likely be effective. This rapid-test methodology accelerates troubleshooting by quickly identifying which types of fixes will be productive.

Conclusion

Radiated emission sources in electronic systems are diverse, interrelated, and often subtle. Clock harmonics, differential-mode current loops, common-mode currents on cables, PCB structures, enclosure apertures, connectors, heat sinks, individual components, and switching power supplies all contribute to the overall emission profile. Understanding these sources enables engineers to design systems with inherently low emissions rather than relying solely on after-the-fact shielding and filtering.

Effective emission control requires attention throughout the design process. Specifying slower clock edges than necessary, selecting low-emission components, designing PCBs with continuous ground planes and minimal loop areas, implementing proper grounding and filtering, and designing enclosures with controlled apertures all contribute to EMC success. Each 6 dB reduction in emissions provides additional margin for compliance and reduces the cost and complexity of shielding and filtering.

As electronic systems become faster and more complex, managing radiated emission sources becomes more challenging. However, the fundamental principles remain consistent: minimize the strength of sources, reduce coupling to radiating structures, and contain fields within shielded boundaries. Mastering these principles enables development of products that comply with regulatory requirements while meeting functional, cost, and schedule targets.