Electronics Guide

Power Line Conducted Interference

Power supply conductors are particularly susceptible to conducted coupling because they connect equipment to shared infrastructure. Interference can propagate in two distinct modes along power lines:

Differential mode (DM): Current flows in opposite directions on the line and neutral conductors, with the interference appearing as a voltage between these conductors. Differential mode interference typically results from switching noise in power converters, rectifier ripple, or modulated loads. At lower frequencies (below approximately 1 MHz), differential mode tends to dominate power line conducted emissions.

Common mode (CM): Current flows in the same direction on both the line and neutral conductors, returning through the ground or parasitic capacitances. Common mode interference often originates from fast-switching circuits with capacitive coupling to the ground plane. At higher frequencies (above approximately 1 MHz), common mode interference typically dominates due to the higher coupling impedances involved.

The distinction between these modes is critical for filter design. Differential mode interference requires series inductors and parallel capacitors between line conductors, while common mode interference requires chokes that present high impedance to signals flowing in the same direction on both conductors, along with capacitors to ground.

Signal Cable Coupling

Signal cables can couple interference through several mechanisms. The cable acts as an antenna that can pick up radiated fields and convert them to conducted signals. Simultaneously, signals within the cable can radiate outward, potentially interfering with other circuits. The effectiveness of this coupling depends on cable length relative to wavelength, termination impedances, and shielding.

Unshielded cables are particularly vulnerable at frequencies where cable length approaches a quarter wavelength or its multiples. At these resonant lengths, the cable efficiently converts between conducted and radiated energy. Even shielded cables can couple interference if shield connections are improperly made or if shield transfer impedance is too high at the frequencies of concern.

Cable routing significantly affects conducted coupling. Cables routed parallel to each other over long distances can couple through mutual capacitance and inductance. Separating sensitive signal cables from power cables and high-current switching circuits reduces this coupling, as does minimizing the length of parallel runs.

Ground Conductor Coupling

Ground conductors, despite their name, are not at zero potential but carry return currents for all circuits sharing the ground system. When high-current circuits share ground conductors with sensitive circuits, the voltage drop along the ground conductor appears as interference in the sensitive circuit. This represents a special case of conducted coupling through shared impedance.

The impedance of ground conductors increases with frequency due to skin effect and inductance, making this coupling more severe for high-frequency signals. A ground conductor only a few centimeters long can present significant impedance at frequencies above 100 MHz, coupling high-frequency noise between circuits that appear to share a common ground.

Near-Field and Far-Field Regions

The characteristics of electromagnetic radiation depend strongly on the distance from the source relative to wavelength:

Near-field region: At distances much less than one wavelength from the source (typically within lambda/2pi), electric and magnetic fields are not necessarily related by the free-space wave impedance. In this region, the field character depends on the source type. Electric dipoles (high-impedance sources like traces with high voltage swings) produce predominantly electric fields, while magnetic dipoles (low-impedance sources like current loops) produce predominantly magnetic fields. Near-field coupling mechanisms are often modeled using capacitive and inductive models rather than full electromagnetic wave analysis.

Far-field region: At distances greater than approximately one wavelength, the electric and magnetic fields become related by the free-space impedance of 377 ohms and propagate together as a unified electromagnetic wave. In this region, the field strength decreases as 1/r (where r is distance), and the wave characteristics become independent of the source type. Far-field coupling is the basis for traditional radio frequency interference.

Most EMC problems within equipment enclosures occur in the near-field region, where the coupling mechanisms can be more complex but also more amenable to localized mitigation. Radiated emissions testing typically measures far-field radiation at distances of 3, 10, or 30 meters.

Antenna Mechanisms

For radiated coupling to occur, both source and victim must have structures that can efficiently convert between circuit currents/voltages and electromagnetic fields. These antenna structures can be intentional (like the antenna in a radio) or unintentional (like a cable or PCB trace).

Common unintentional antenna structures include:

• Wire or trace monopoles: A conductor connected at one end acts as a quarter-wave monopole antenna at frequencies where its length equals one-quarter wavelength. A 7.5 cm trace resonates at 1 GHz.
• Loop antennas: A current-carrying loop radiates as a magnetic dipole. PCB signal loops and cable loops between signal and return paths act as loop antennas.
• Slot antennas: Gaps or slots in shielding enclosures radiate when current must flow around the slot. A slot resonates when its length equals one-half wavelength.
• Horn antennas: Openings in enclosures where cables enter can act as small horn antennas, launching radiation into free space.

The efficiency of unintentional antennas varies dramatically with frequency. Below the first resonance, radiation efficiency increases as f-squared, making even short structures effective radiators at sufficiently high frequencies.

Field-to-Cable Coupling

External electromagnetic fields can couple to cables through two mechanisms:

Electric field coupling: The electric field component induces charges and hence voltages on the cable conductors. This coupling is proportional to the cable-to-ground capacitance and is most effective when the cable is electrically short (much less than a wavelength). High-impedance circuits are more susceptible to electric field coupling.

Magnetic field coupling: The magnetic field component induces a voltage in any loop formed by the cable and its return path, according to Faraday's law. The induced voltage is proportional to the loop area and the rate of change of the magnetic field. Low-impedance circuits with large loop areas are more susceptible to magnetic field coupling.

Shielded cables reduce field-to-cable coupling by intercepting the fields before they reach the inner conductors. Shield effectiveness depends on shield coverage, the quality of shield terminations, and the shield transfer impedance at the frequencies of concern.

Fundamentals of Capacitive Coupling

The capacitance between two conductors depends on their geometry, separation, and the dielectric material between them. For parallel conductors of length L, radius r, and center-to-center spacing d:

C = (pi * epsilon * L) / ln(d/r)

For parallel PCB traces of width w, separation s, and length L over a ground plane:

C approximately equals (epsilon * L * h) / s

where h is the trace height above the ground plane.

The current coupled into the victim circuit through this capacitance is:

I = C * (dV/dt) = j * omega * C * V

This shows that capacitive coupling increases linearly with frequency and with the rate of voltage change. Fast-edged digital signals with high dV/dt are particularly effective at coupling capacitively to adjacent circuits.

Capacitive Coupling in PCB Design

On printed circuit boards, capacitive coupling occurs between adjacent traces, between traces on different layers, and between components. The coupling is strongest when:

• Traces run parallel for long distances
• Trace separation is small relative to the trace-to-ground spacing
• The victim circuit has high impedance
• The source signal has fast edges or high frequency content

Reducing capacitive coupling on PCBs involves increasing trace separation, reducing parallel run lengths, using ground traces or planes between sensitive signals, and keeping trace heights low relative to separation. For especially sensitive signals, guard traces connected to ground can provide electrostatic shielding.

Layer-to-layer capacitive coupling can be significant in multilayer boards. A signal trace on one layer couples capacitively to traces on adjacent layers, particularly where they cross. This is usually less problematic than parallel coupling because the overlap area is small, but it can become significant at very high frequencies or in dense layouts.

Capacitive Coupling Between Cables

Cables routed in parallel develop mutual capacitance that allows electric field coupling. This coupling is strongest for unshielded cables and increases with cable length and proximity. In cable bundles, the outer cables shield inner cables somewhat, but significant coupling can still occur.

The capacitance per unit length between parallel cables depends on their diameter and separation. Typical values range from a few picofarads per meter for well-separated cables to tens of picofarads per meter for cables in contact. Over a 1-meter cable run, this can represent capacitances of 10-50 pF, which becomes a low-impedance path at frequencies above 100 MHz.

Shielded cables reduce capacitive coupling by containing the electric field within the shield. However, the shield must be properly grounded; a floating shield can actually increase coupling by acting as an intermediate conductor.

Component-Level Capacitive Coupling

Within components and between adjacent components, capacitive coupling can cause unexpected interactions:

• Package parasitic capacitance: Component leads and internal bonding wires have capacitance to each other and to the package. These parasitics can couple high-frequency signals between nominally independent pins.
• Heatsink coupling: Heatsinks attached to switching devices can couple switching noise to other circuits through their capacitance. Insulating thermal interface materials reduce but do not eliminate this coupling.
• Inter-component coupling: Components mounted close together on a PCB can couple through the electric field. This is especially problematic for high-gain amplifiers near digital logic.

Mutual Inductance Fundamentals

The mutual inductance M between two circuits quantifies their magnetic coupling. When current i1 flows in circuit 1, it induces a voltage in circuit 2:

V2 = M * (di1/dt) = j * omega * M * I1

Mutual inductance depends on the geometry of both circuits and their relative orientation. For two parallel conductors of length L and separation d carrying return currents at distances d1 and d2:

M = (mu * L / 2 * pi) * ln[(d + d2) / d1]

The coupling coefficient k relates mutual inductance to self-inductances:

k = M / sqrt(L1 * L2)

For loosely coupled circuits typical in EMC scenarios, k is usually much less than 0.1, but even small coupling can cause problems when source currents are large or victim circuits are sensitive.

Loop Area and Orientation Effects

The voltage induced in a victim circuit depends critically on the area of the loop formed by the signal and its return path. Reducing loop area is one of the most effective ways to reduce inductive coupling:

• Keeping signal traces close to their return paths (ground plane or return trace)
• Using differential signaling where the two conductors carry equal and opposite currents
• Twisting wire pairs so that induced voltages in successive twists cancel
• Using coaxial cables where the shield carries the return current coaxially with the signal

The orientation of the victim loop relative to the source field also matters. Maximum coupling occurs when the loop plane is perpendicular to the magnetic field (loop axis parallel to field). No coupling occurs when the loop plane is parallel to the field (loop axis perpendicular to field). In practice, random orientations result in coupling somewhere between these extremes.

Inductive Coupling in PCB Design

On printed circuit boards, inductive coupling occurs primarily between signal loops and is often called crosstalk along with capacitive coupling. The magnetic field from current in one trace induces voltage in adjacent traces and their return paths.

With a solid ground plane beneath signal traces, the return current tends to flow directly beneath the signal trace, minimizing loop area. This tight coupling between signal and return reduces both the source's ability to generate external magnetic fields and the victim's ability to pick them up.

Disruptions in the ground plane, such as splits or gaps, force return current to take longer paths, increasing loop area and magnetic coupling. Ground plane slots should never cross beneath sensitive high-speed signal traces.

For differential signals, inductive coupling is dramatically reduced because the currents in the two conductors flow in opposite directions, producing magnetic fields that largely cancel. This is one of the key advantages of differential signaling for EMC.

Transformer Coupling

When source and victim circuits share a magnetic core (intentionally or accidentally), coupling is greatly enhanced. Transformer coupling can occur in:

• Multi-winding transformers: Noise on one winding couples to others through the shared magnetic circuit. Inter-winding capacitance provides additional coupling paths.
• Shared magnetic cores: Multiple inductors wound on the same core (as in some multi-phase power supplies) can couple through the common flux path.
• PCB via magnetic coupling: Vias carrying high-current signals generate magnetic fields that can link with nearby via structures.

Transformer coupling is usually addressed through careful winding arrangement, shielding windings, or physical separation of magnetically sensitive circuits from magnetic components.

Ground Impedance Coupling

The most common form of common-impedance coupling involves shared ground conductors. When a high-current circuit and a sensitive circuit share a ground path, current from the high-current circuit creates a voltage drop along the ground impedance that appears as interference in the sensitive circuit.

Consider two circuits sharing a ground wire of length L and inductance approximately 10 nH/cm. If circuit A draws a current changing at 100 A/microsecond (typical for digital logic), the voltage developed across 10 cm of ground wire is:

V = L * (di/dt) = 100 nH * 100 A/microsecond = 10 V

This 10 V appears in series with circuit B's signal, potentially overwhelming it. Even at DC or low frequencies, resistance creates coupling: a 10 milliohm ground path carrying 10 A creates a 100 mV offset.

Solving ground impedance coupling requires either reducing the shared impedance (using heavier conductors, shorter paths, or ground planes) or eliminating the shared path (using separate ground returns for different circuits, or single-point grounding).

Power Supply Impedance Coupling

Power supply lines also represent shared impedances. Variations in current draw by one circuit cause voltage variations at the power supply that affect all other circuits powered from the same source.

The power supply's output impedance increases with frequency due to the inductance of output capacitors and connecting traces. Even a low-impedance power supply can show significant impedance at the frequencies where digital circuits generate noise (typically 10 MHz to 1 GHz).

Decoupling capacitors address power supply impedance coupling by providing a local, low-impedance source for high-frequency current demands. The capacitor's impedance must be low at the frequencies of concern, which requires attention to the capacitor's equivalent series inductance (ESL) and the inductance of its connection to the circuit.

Multiple stages of decoupling are often used: bulk capacitors (10-100 microfarads) for low-frequency decoupling, ceramic capacitors (0.1-10 microfarads) for mid-frequency decoupling, and small ceramic or integrated capacitors (10-100 nF) for high-frequency decoupling close to IC power pins.

Signal Return Path Coupling

Signal circuits can also exhibit common-impedance coupling when return currents share paths. This is particularly problematic in systems with star grounding, where return currents from different circuits must flow through common segments of the ground structure.

In high-speed digital systems, return currents naturally flow in the path directly beneath the signal trace on the ground plane. If this path is interrupted (by slots, vias, or plane changes), return current must find an alternate route, potentially sharing paths with other signals and creating common-impedance coupling.

Proper return path management ensures that each signal has a continuous, low-impedance return path that does not intersect with returns from other signals. This is a fundamental principle of high-speed PCB design and EMC.

Star vs. Multi-Point Grounding

Two fundamental grounding strategies address common-impedance coupling differently:

Star (single-point) grounding: All circuit returns connect to a single point, eliminating shared ground paths. This approach is effective at low frequencies (below approximately 100 kHz) where ground conductor inductance is negligible. At higher frequencies, the inductance of the long ground conductors required for star grounding can create problems.

Multi-point grounding: Circuits connect to a low-impedance ground plane at multiple points. This minimizes ground inductance at high frequencies but allows ground current from different circuits to mix in the plane. At low frequencies, resistive drops in the plane can cause coupling.

Many systems use hybrid grounding: star grounding for low-frequency analog circuits and multi-point grounding for high-frequency digital circuits. The transition frequency depends on circuit impedances and acceptable coupling levels, but is typically in the range of 10-100 kHz.

Slot Antenna Theory

A slot in a conductive plane is the electromagnetic dual of a dipole antenna. By Babinet's principle, the radiation pattern of a slot is the same as that of a dipole of the same dimensions, rotated 90 degrees in polarization.

A slot radiates most efficiently when its length is approximately one-half wavelength. At this resonant length:

L = lambda/2 = c / (2f)

A 15 cm slot resonates at 1 GHz, while a 3 cm slot resonates at 5 GHz. Even slots much shorter than a half wavelength can couple significant energy at high frequencies.

The shielding effectiveness of an enclosure with a slot is limited by:

SE approximately equals 20 * log(lambda / 2L) dB

for slot length L much less than wavelength. This shows that shielding effectiveness decreases 20 dB per decade of frequency increase, making slots increasingly problematic at higher frequencies.

Enclosure Aperture Design

Practical enclosures require openings for ventilation, displays, controls, and cable entry. Each opening potentially compromises shielding effectiveness:

Ventilation holes: Arrays of small holes provide better shielding than a single large opening of the same total area. The critical dimension is the longest dimension of each individual hole, not the total open area. Honeycomb panels with thousands of small cells provide excellent shielding while allowing substantial airflow.

Seams and joints: Where enclosure panels meet, the electrical continuity across the joint determines shielding effectiveness. Gaps at seams act as slots. Gaskets, finger stock, and overlapping joints maintain conductivity and reduce slot length.

Cable penetrations: Cables entering an enclosure must be filtered or shielded to prevent them from acting as antennas that couple energy across the shield boundary. Feedthrough filters provide both shielding and filtering at the enclosure boundary.

Display windows: Non-conductive display windows create large apertures. Conductive coatings or embedded wire mesh maintain shielding while allowing visibility. The mesh size must be small compared to wavelength at the frequencies of concern.

PCB Ground Plane Slots

Slots in PCB ground planes create similar coupling problems at the board level. A ground plane slot forces return current to flow around the slot, increasing loop area and potentially creating a slot antenna structure.

Common causes of ground plane slots include:

• Routing traces on inner layers that cut through ground planes
• Clearances around through-hole component leads
• Split ground planes for analog/digital isolation
• Thermal relief patterns around ground vias

When a high-speed signal trace crosses a ground plane slot, the return current must detour around the slot, creating a large loop that both radiates and picks up interference. This situation should be avoided by careful layer stack planning and routing discipline.

If ground plane slots are unavoidable (as in some analog/digital partition schemes), signals should cross only at a bridging point where a low-impedance path for return current is provided, typically by decoupling capacitors or a narrow ground connection.

Coupled Transmission Line Model

Two parallel cables can be modeled as coupled transmission lines with both self-parameters (characteristic impedance, propagation constant) and mutual parameters (mutual capacitance, mutual inductance).

For a cable pair, the coupling can be characterized by near-end crosstalk (NEXT) and far-end crosstalk (FEXT):

Near-end crosstalk: Noise appearing at the end of the victim cable nearest to the source signal input. NEXT results from the combination of capacitive and inductive coupling, which add in the backward direction.

Far-end crosstalk: Noise appearing at the end of the victim cable farthest from the source signal input. FEXT results from the difference between capacitive and inductive coupling, which partially cancel in the forward direction.

For typical cable geometries in air or low-permittivity insulators, capacitive and inductive coupling coefficients are approximately equal, resulting in significant NEXT and relatively low FEXT. This is why crosstalk is often more problematic near the source end of a cable run.

Cable Separation Requirements

Coupling between cables decreases with increasing separation, but the relationship is logarithmic rather than linear for both capacitance and inductance. Doubling the separation reduces coupling by only about 6 dB (factor of 2).

Practical cable separation guidelines depend on cable types and signal characteristics:

• Power cables and sensitive signals: Separate by at least 10 cm where possible; use shielded cables if closer spacing is required
• High-speed digital and analog signals: Separate by at least 5 cm or use shielded cables
• Similar signal types: Closer spacing acceptable but minimize parallel run length

When physical separation is impossible, other techniques reduce coupling: shielding (especially if shields are properly grounded at both ends), twisting (which reduces the effective loop area for magnetic coupling), and routing at right angles (which minimizes coupling length).

Shield Effectiveness for Cable Coupling

Shielded cables reduce cable-to-cable coupling by containing fields within the shield. Shield effectiveness depends on:

Shield coverage: Braided shields typically provide 85-95% coverage, while solid tube shields provide 100% coverage. Gaps in coverage allow field leakage.

Shield material: Higher conductivity materials provide better shielding. Copper braid is common; aluminum foil with drain wire is lighter but may have lower effectiveness at higher frequencies.

Shield termination: The shield must make a 360-degree connection to the enclosure or connector shell for maximum effectiveness. Pigtail connections (shield wire connection) add inductance that degrades high-frequency shielding.

Transfer impedance: This parameter characterizes how much voltage appears on the inner conductor due to current flowing on the shield. Lower transfer impedance indicates better shielding. Transfer impedance typically increases with frequency due to shield inductance and skin effect.

Capacitive-Inductive Combined Coupling

Crosstalk between parallel conductors always involves both capacitive and inductive coupling. These mechanisms have different characteristics that affect the total coupling:

• Capacitive coupling produces a victim signal proportional to dV/dt (voltage rate of change)
• Inductive coupling produces a victim signal proportional to dI/dt (current rate of change)
• In near-end crosstalk, both mechanisms produce signals in the same direction, adding constructively
• In far-end crosstalk, the mechanisms produce signals in opposite directions, partially canceling

The relative contribution of each mechanism depends on the source and victim impedances. High-impedance circuits are more susceptible to capacitive coupling, while low-impedance circuits are more susceptible to inductive coupling.

Differential signaling exploits this interaction: by carrying equal and opposite signals on two conductors, both capacitive and inductive coupling from external sources tend to appear as common-mode signals that are rejected by the differential receiver.

Conducted-Radiated Hybrid Coupling

Cables attached to equipment can act as antennas, converting conducted interference to radiated emissions and vice versa. This creates hybrid coupling paths that combine conducted and radiated mechanisms:

Conducted to radiated: Interference conducted on a cable causes current to flow on the cable, which radiates. The cable acts as an unintentional antenna with efficiency depending on cable length relative to wavelength.

Radiated to conducted: External electromagnetic fields induce currents on cables, which then appear as conducted interference on connected circuits. The cable acts as a receiving antenna.

These hybrid mechanisms are why radiated emissions testing often reveals problems traceable to poor filtering on I/O cables, and why susceptibility testing may show failures related to cable common-mode currents.

Common-mode chokes on cables address these hybrid paths by presenting high impedance to the common-mode (antenna mode) currents while allowing differential (signal mode) currents to pass. Ferrite clamps around cables are a simple form of common-mode choke.

Common-Impedance with Field Coupling

Common-impedance coupling and field coupling can interact when shared ground structures serve as both return current paths and field shields:

A ground plane simultaneously carries return currents (creating common-impedance coupling opportunities) and shields against field coupling. If the plane is incomplete or has high impedance, both mechanisms are affected. Slots in a ground plane increase common-impedance coupling (by lengthening return paths) and enable field coupling (by creating slot antennas).

Enclosure grounding involves similar interactions. The enclosure shield intercepts external fields, but the resulting currents must flow through the ground system, potentially coupling to internal circuits through shared ground impedances.

Low-Frequency Dominated Mechanisms

Some coupling mechanisms are most effective at lower frequencies and decrease in significance as frequency increases:

Resistive common-impedance coupling: The DC and low-frequency coupling through shared resistance dominates at frequencies where the resistive drop exceeds the inductive drop. As frequency increases, the inductive impedance (proportional to frequency) eventually exceeds the resistance.

Conductive leakage: Leakage through imperfect insulation represents a resistive coupling path that is relatively frequency-independent, making it most significant at low frequencies where other coupling mechanisms are weak.

High-Frequency Dominated Mechanisms

Most coupling mechanisms become more significant at higher frequencies:

Capacitive coupling: Current through a capacitance is proportional to frequency (I = omega*C*V), so capacitive coupling increases at 20 dB/decade.

Inductive coupling: Voltage induced by mutual inductance is proportional to frequency (V = omega*M*I), so inductive coupling also increases at 20 dB/decade.

Radiated coupling: Antenna efficiency generally increases with frequency (as electrical size increases), and the energy content of fields at a given amplitude increases with frequency.

Inductive common-impedance: Ground inductance impedance is proportional to frequency, making this coupling mechanism increase at 20 dB/decade.

Resonance Effects

Many structures exhibit resonances where coupling dramatically increases:

Cable resonances: Cables resonate when their length is a multiple of quarter wavelength. At resonance, the cable acts as an efficient antenna and voltage/current standing waves develop, potentially creating very high voltages or currents at certain points.

Enclosure resonances: Shielded enclosures act as cavity resonators at frequencies where dimensions are comparable to half wavelength. At cavity resonance, fields inside the enclosure can be much higher than external fields, even for well-shielded enclosures.

Ground plane resonances: PCB ground planes exhibit parallel-plate resonances between power and ground planes, and patch antenna resonances at edges. These resonances can greatly increase both emissions and susceptibility at specific frequencies.

Identifying and managing resonances is critical for EMC. Damping resistors, ferrite loading, and careful dimensioning can shift or suppress problematic resonances.

Wideband vs. Narrowband Coupling

The frequency dependence of coupling combines with the frequency content of interference sources to determine the overall coupling spectrum:

Narrowband sources: Intentional emissions like radio transmitters or clock oscillators concentrate energy at specific frequencies. Coupling at these frequencies determines the interference level. Narrowband interference can often be addressed by filtering at the specific frequencies involved.

Wideband sources: Broadband sources like digital logic, switching converters, or electrostatic discharge contain energy across a wide frequency range. The coupling spectrum combines the source spectrum with the frequency response of the coupling mechanism. Wideband interference is harder to filter and often requires broadband suppression techniques.

When analyzing EMC problems, it is important to consider the source spectrum, the coupling frequency response, and any resonances that might enhance coupling at specific frequencies. The worst-case interference often occurs when a strong spectral component from the source coincides with a resonance in the coupling path.

Coupling Transfer Functions

Coupling between circuits can be characterized as a transfer function relating victim response to source excitation. For linear coupling mechanisms:

V_victim(f) = H_coupling(f) * V_source(f)

The coupling transfer function H(f) captures all the frequency-dependent effects of the coupling path. It can be measured using a network analyzer or calculated from circuit models.

For cable coupling, the transfer function is often expressed as crosstalk ratio (in dB):

Crosstalk = 20 * log(V_victim / V_source) dB

Typical crosstalk requirements range from -30 dB for general applications to -60 dB or better for sensitive analog circuits.

Field Probing Techniques

Near-field probes can directly measure the electric and magnetic fields responsible for radiated and capacitive/inductive coupling:

E-field probes: Small monopole antennas or active probes with high-impedance inputs sense the electric field component. They are useful for identifying sources of capacitive coupling and areas of high voltage.

H-field probes: Small loop antennas sense the magnetic field component. They are useful for identifying sources of inductive coupling and locating high-current paths. Shielded loops reject electric field pickup for cleaner magnetic field measurements.

By scanning near-field probes over a PCB or system, coupling sources can be localized and the effectiveness of mitigation measures can be evaluated.

Time-Domain Techniques

Time-domain measurements reveal coupling behavior that may not be apparent from frequency-domain analysis:

Time-domain reflectometry (TDR): A fast pulse launched into a transmission line reflects from impedance discontinuities. The timing and magnitude of reflections reveal the location and severity of coupling-related impedance variations.

Time-domain crosstalk measurement: Observing the victim response to a pulse on the aggressor directly shows the coupling waveform, including both near-end and far-end components. This is particularly useful for digital signal integrity analysis.

Transient capture: Capturing interference events triggered by specific system operations (such as motor starts, relay operations, or ESD events) reveals coupling paths that are only active during specific conditions.

Source Suppression

Reducing the interference at its source is often the most effective approach:

• Slowing edge rates on digital signals reduces high-frequency content
• Filtering at the source prevents noise from reaching coupling paths
• Reducing voltage and current magnitudes directly reduces coupling levels
• Proper circuit layout minimizes loop areas and antenna structures

Path Attenuation

Reducing coupling along the path from source to victim:

• Physical separation reduces both field coupling and mutual parameters
• Shielding intercepts fields before they can couple
• Filtering on cables attenuates conducted coupling
• Proper grounding and return path management reduces common-impedance coupling

Victim Hardening

Making the victim circuit less susceptible to interference:

• Reducing victim loop area decreases field pickup
• Using differential signaling provides common-mode rejection
• Filtering at the victim input attenuates out-of-band interference
• Increasing signal levels improves signal-to-interference ratio